RF amplifier

ABSTRACT

A RF amplifier device ( 22 ) including an amplifier element ( 24 ) compensated by a compensating circuit ( 26, 28 ) with respect to its output capacitance and frequency decoupled from its power supply ( 26 ), wherein the decoupling circuit is directly connected to the compensating circuit ( 26, 28 ) and a RF amplifier device including an amplifier element ( 56, 80 ) and a compensating circuit comprising an internal shunt inductor having a compensating inductance ( 58, 60, 62 ) and a compensating capacitance ( 64, 92 ) and arranged in parallel to a terminal of the amplifier element ( 56, 80 ) to compensate a terminal capacitance of the amplifier element ( 56, 80 ), and a decoupling and power supply lead ( 76, 98 ) which is connected to the compensating capacitance ( 64, 92 ) and/or a decoupling circuit ( 100 ) and/or a combination of the compensating capacitance and the decoupling circuit ( 130 ) and a module thereof and a method for decoupling the mentioned RF amplifier device.

The invention relates to a RF amplifier device comprising an amplifierelement with a frequency dependent gain, said frequency dependence beingcaused by an input and/or an output capacitance, said frequencydependence being compensated by a compensating circuit.

Telecommunication providers place transmitters or base stationsthroughout the landscape, so that everybody can use her or his telephoneat any place. These base stations comprise amplifiers. These amplifiersamplify digital signals which are modulated on a high frequency carrier(1 or 2 GHz). The result is a very complex signal having a complexspectrum. For instance GSM, edge-GSM and CDMA are standards which areused for transmitting the data between the base stations and the mobilephones. In successors of these systems, a wide band-CDMA signal, a socalled W-CDMA, is used. As the transmitted amount of data is very large,the processing of the data in W-CDMA is the most complex one.

If two (or more) signals with different frequencies are amplified,differential tones or frequencies occur. For example, a first signalwith a first frequency and a second signal with a second frequency, willresult in a differential third signal C with a third frequency thatequals the first frequency minus the second frequency. The differentialthird signal is not desired since it generates bias modulation effects,resulting in poor linearity and spectrum asymmetry of the amplifier(memory effects). In order to avoid such unwanted effects, it isnecessary to eliminate the differential third signal. Therefore, thebase station amplifiers require broad band decoupling circuits fordifferential tones. For standards like (multi carrier) W-CDMA decouplingis required up to 50 MHz. Lower frequencies can be shorted “far away”from the transistor die with electrolytic capacitances, but frequenciesabove 5–10 MHz need a very short path to the decoupling capacitances inorder to be shorted sufficiently.

Furthermore, in conventional base station amplifiers the supply voltageis connected to the amplifying transistor via a ¼ wavelength line (λ/4line). The ¼ wavelength line technique is space-consuming and providesonly a narrow band solution. The ¼ wavelength line is based on thetheory that, if a terminal B of a two-port network is short-circuited,which means a voltage equal to zero and a short circuit currentI_(short), then an open circuit situation is found at a terminal A ofthe two-port network, which means a current equal to zero and a shortcircuit Voltage V_(short) is measured on terminal A, and vice versa. The¼ wavelength line can thus be used as a filter. However, due to the pathlength of the ¼ wavelength line it is very difficult to obtain alow-ohmic short for higher frequencies up to 10 MHz. Furthermore, the ¼wavelength line technology is space consuming which is not desired in aworld of miniaturization.

FIG. 1 shows the conventional bias voltage supply arrangement comprisinga power supply connection layer 16 and decoupling capacitances 18. A RFamplifier device 2 comprises an active die 4 and a DC blockingcapacitance 6. The active die 4 is connected via bond wires 8 forming anINSHIN inductance to the DC blocking capacitance 6 and via further bondwires 9 to a matching circuit 10. The matching circuit 10 is connectedto a RF-short capacitance 20 and to the decoupling capacitances 18 via a¼ wavelength line 12. The decoupling capacitances 18 are connectedbetween the power supply connection layer 16 and ground 14.

In general, an amplifier of a base station features an INSHIN inductor(INSHIN=INternal SHunt INductor) at its output in order to compensatethe output capacitance. The INSHIN inductor is incorporated in thetransistor package and is formed by the bond wires 8 which are connectedwith the active die 4 and the DC blocking capacitance 6. The INSHINinductor is commonly used in RF-power transistors having an output powerabove 20 W. The INSHIN inductor which is formed by the bond wires 8between the active die 4 and the DC blocking capacitance 6, is in serieswith a DC blocking capacitance 6. This serial circuit is in parallel tothe output of the active die 4. This INSHIN inductor 8 is connected toground with the DC blocking capacitance 6.

As mentioned above in connection with the principle of ¼ wavelengthline, the short of a ¼ wavelength line can be compared with aRF-termination capacitance which is connected to ground, which blocksthe DC and shorts the RF-signals. When frequency changes, then theelectrical length is not ¼ wavelength anymore, and the DC bias of thepower supply connection layer 16 will interfere with the matchingcircuit 10 so the ¼ wavelength line 12 works only over a rather narrowfrequency band or, with other words, it is narrow-banded. The electricallength of a ¼ wavelength line stub on 2 GHz is 7–20 mm which depends onthe material of the printed circuit board.

The ¼ wavelength line solution introduces a long path from thetransistor to the capacitances. By increasing the path length increasedthe impedance is increased also. For very low frequencies (<500 kHz) theimpedance of the path length is negligible, but the higher thedifferential frequency is the more important the path length becomes.This makes it difficult to achieve a good short on the actual transistorat frequencies above 5–10 MHz.

For example, if a capacitance for shorting the differential frequency isapproximately 20 mm spaced apart from the nearest transistor die, thisproves to be too long a distance to short differential frequencies whichare higher than 5 MHz. In other words, if the DC bias is provided, thena ¼ wavelength line is required because the DC bias of the power supplyconnection layer 16 must not disturb the functioning of the RF matchingcircuit 10. This solution consumes space on circuit boards. In a worldof extremely small designs it is undesired to consume too much space oncircuit boards. Also the path length from the transistor to thecapacitances which shorts the differential frequency, is undesirablylong.

For a good performance of the amplifier the differential frequency mustbe shorted. Because influences from the power supply to the matchingcircuit have to be avoided, decoupling capacitances 18 are provided toshort the differential frequencies at lower frequencies. The decouplingcapacitances 18 are connected to the power supply connection layer 16and are required as additional filters.

In general, an amplifier of a base station features a shunt inductor 8at its output in order to compensate the output capacitance. The shuntinductor is commonly used with RF-power transistors with an output powerabove 20 W. The shunt inductor 8 is connected in series with a DCblocking capacitance 6 and is coupled to ground through the DC blockingcapacitance 6.

European patent application EP 0 368 329 A discloses a self equalizingmulti-stage radio frequency power amplifier. The linearity andefficiency of a radio frequency two-stage power amplification device areincreased by employing two tuned circuits in a driver stage and twotuned circuits in a high power stage. After selecting elements of tunedcircuits in order to optimize the high power stage for efficiency,linearity and power output, elements are selected for tuned circuits inorder to cause the intermodulation output components associated with thedriver stage to have a 180° phase angle relative to the intermodulationoutput components associated with the individual high power stage. Thisphase angle relationship has the effect that the intermodulation outputcomponent products from the driver and high power stages are canceled.The circuit comprising the shunt inductor is in series with a DCblocking capacitance is connected to ground and forms a circuit stageoutside of the transistor device.

British patent application GB 2 225 683 A discloses a high frequencyamplifier which prevents parametric oscillations. Parasitic, parametricoscillations in the amplifier may be prevented by coupling a resonantcircuit to either the input or the output of the amplifier. The resonantcircuit has a resonant frequency equal to one half the frequency of theamplified signal. The resonant circuit comprises only a portion of theinductor connected to the output or input of the amplifier.

In patent application GB 2 225 683 A the DC power supply is not fed to aDC blocking capacitance. Furthermore, the differential frequencies arenot decoupled by a DC blocking capacitance. Also the shunt inductance isa device separate from the transistor and is not accessible for thecustomer.

It is an object of the present invention to provide a an RF amplifierdevice for decoupling at higher frequencies.

To achieve this object, the present invention provides a RF amplifierdevice according to the opening paragraph, said compensating circuitbeing coupled to a power supply terminal, said power supply terminalbeing connected to a decoupling circuit.

In the device according to the invention the path length from thedecoupling circuit to the amplifying transistor is dramatically reducedby connecting the power supply and/or the decoupling circuit to thecompensating capacitance. This results in a higher decoupling frequencyand a simple bias connection. Furthermore the area required by thecircuit is decreased, resulting in lower production costs.

According to a preferred embodiment of the device according to theinvention the RF amplifier device comprises an amplifier element and acompensating circuit comprising an internal shunt inductor having acompensating inductance and a compensating capacitance, saidcompensating circuit being arranged in parallel to the amplifier elementoutput to compensate an input and/or an output capacitance of thedevice, said device further comprising a power supply terminal connectedto said amplifier element through said decoupling circuit, whereby thedecoupling circuit is coupled to the compensating capacitance via saidpower supply terminal. By coupling the power supply and the decouplingcircuit to the compensating capacitance, the path length from thedecoupling circuit to the amplifying transistor is reduced. The resultsare a higher decoupling frequency and a simple bias connection.

According to a further preferred embodiment of the device according tothe invention the decoupling circuit is coupled to the compensatingcapacitance through an inductance element. By using the connectingelement as a functional element, space of the circuit board and time andmoney in production can be saved because it is not necessary to buy andmount a separate inductance.

According to another further preferred embodiment of the deviceaccording to the invention the inductance element is at least one bondwire. The value of the inductance can be easily matched to theconditions of the customer by the number of bond wires, the diameter andthe length thereof.

According to another further preferred embodiment of the deviceaccording to the invention the decoupling circuit is connected betweenground and the power supply. The result is a simpler arrangement.Furthermore, the space required by the circuit is decreased.

According to another further preferred embodiment of the deviceaccording to the invention the decoupling circuit comprises at least onedecoupling capacitance. The impedance of the decoupling capacitance canbe matched to the conditions of the circuit by the number of thedecoupling capacitances.

According to another further preferred embodiment of the deviceaccording to the invention a power supply line is provided comprising apower supply connection area, a decoupling circuit connection area and abond wire connection area, which line is arranged next to thecompensating capacitance. The result is a much higher decouplingfrequency and a simple power supply connection layer ion.

According to another preferred embodiment of the device according to theinvention said RF amplifier device is a transistor.

According to another further preferred embodiment of the deviceaccording to the invention, the amplifier element with the compensatingcircuit, the decoupling circuit and the connection line having theconnection area for a power supply and a connection area for thedecoupling circuit are arranged on a circuit board, wherein theconnection line is located on the circuit board next to the compensatingcapacitance. The narrow location of the above mentioned functional partscontribute to decrease the space of the circuit.

According to another further preferred embodiment of the deviceaccording to the invention a RF amplifier device including an amplifierelement and a compensating circuit is disclosed, comprising an internalshunt inductor having a compensating inductance in series with acompensating capacitance, which are arranged in parallel to a terminalof the amplifier element to compensate a terminal capacitance of theamplifier element, said device further comprising a decoupling and powersupply lead connected to the compensating capacitance and/or adecoupling circuit and/or a combination of the compensating capacitanceand the decoupling circuit.

In the device the path length from the decoupling circuit to theamplifying transistor is reduced by connecting the power supply and/orthe decoupling circuit to the compensating capacitance. This leads to ahigher decoupling frequency and a simple bias connection. Furthermorethe space required by the circuit is decreased, resulting in a decreasein production costs.

According to another preferred embodiment of the device according to theinvention, the terminal of the amplifier element is an input terminaland/or an output terminal of the amplifier element. An advantageousfeature of this embodiment is that the capacitance of the input and/oroutput terminal are compensated.

According to another further preferred embodiment of the deviceaccording to the invention, the decoupling circuit is connected to thecompensating capacitance through an inductance element.

According to another further preferred embodiment of the deviceaccording to the invention, the inductance element comprises at leastone bond wire.

According to another further preferred embodiment of the deviceaccording to the invention, the decoupling circuit and/or thecombination of the compensating capacitance and the decoupling circuitare/is connected between the decoupling and power supply lead and thecompensating capacitance or between the decoupling and power supply leadand the terminal of the amplifier element. In this preferred embodimentthe path length from the decoupling circuit to the amplifying transistoris reduced by connecting the power supply and/or the decoupling circuitto the compensating capacitance. This leads to a higher decouplingfrequency and a simple bias connection. Furthermore the space requiredby the circuit is decreased and the decoupling frequency is increased.

According to another further preferred embodiment of the deviceaccording to the invention, the decoupling circuit comprises at leastone decoupling capacitance.

According to another further preferred embodiment of the deviceaccording to the invention, the amplifier element is a transistor.

According to another further preferred embodiment of the deviceaccording to the invention, the amplifier element with the compensatingcircuit and/or the decoupling circuit and/or the combination of thecompensating capacitance and the decoupling circuit and the decouplingand power supply lead are arranged on a circuit board. The integrationon one circuit board leads to a further miniaturization of the circuit,which saves space on a circuit board.

A module according to the invention comprises a RF amplifier device,said module further comprising a mounting base for a discrete transistoron which a printed circuit board (pcb) is soldered; a matching network;a bias circuit; at least one decoupling capacitance. The module has theadvantage, that the path lengths are decreased. This leads to a higherdecoupling frequency.

According to a preferred embodiment of the module according to theinvention, the printed circuit board is a multilayer printed circuitboard. This feature decreases the area of the circuit boardtremendously.

According to a further preferred embodiment of the module according tothe invention, the printed circuit board contains all or a part of thematching network and/or the bias circuit. This feature enhances theflexibility of circuit design and mounting technology, because thematching network and/or the bias circuit can be mounted at least to apart on the printed circuit board.

According to another further preferred embodiment of the moduleaccording to the invention, a signal path is on a top layer and adecoupling and power supply path is on a middle layer of the pcb or viceversa.

According to another further preferred embodiment of the moduleaccording to the invention, the decoupling and power supply path is inparallel to the dies of an amplifier element, a compensatingcapacitance, a decoupling circuit, a combination of the compensatingcapacitance and the decoupling circuit.

A method according to the invention is arranged for decoupling a RFamplifier device comprising an amplifier element with a frequencydependent gain, said frequency dependence being caused by an inputand/or an output capacitance, said frequency dependence beingcompensated by a compensating circuit, said compensating circuit beingcoupled to a power supply terminal, said power supply terminal beingcoupled to ground via a frequency dependent impedance.

In the method according to the invention the path length from thedecoupling circuit to the amplifying transistor is dramatically reducedby connecting the power supply and/or the decoupling circuit to thecompensating capacitance. This results in a higher decoupling frequencyand a simple bias connection. Furthermore the area required by thecircuit is decreased, resulting in lower production costs.

These and various other advantages and features of novelty whichcharacterize the present invention are pointed out with particularity inthe claims annexed hereto and forming a part hereof. However, for abetter understanding of the invention, its advantages, and the objectobtained by its use, reference should be made to the drawings which forma further part hereof, and to the accompanying descriptive matter inwhich there are illustrated and described preferred embodiments of thepresent invention.

FIG. 1 shows a conventional arrangement of the bias supply and thedecoupling capacitances;

FIG. 2 shows an arrangement corresponding to the present invention; and

FIG. 3 is a diagram showing the difference in magnitude of the impedancebetween an RF amplifier device with traditional decoupling and an RFamplifier device with a decoupling arrangement of a preferred embodimentof the invention;

FIG. 4 shows a standard case of a RF discrete power transistor;

FIG. 5 shows the case of an adapted RF power transistor with two extraleads;

FIGS. 6–9 show different embodiments of INSHIN decoupling and powersupply in discrete transistors;

FIGS. 10–13 show different embodiments in INSHIN decoupling and powersupply in modules.

FIG. 2 shows an arrangement according to the present invention. TheFigure shows an RF amplifier device 22 which comprises the active diewith a semiconductor amplifier element 24 and the DC blockingcapacitance 26. The active die 24 is connected via bond wires 28 withthe DC blocking capacitance 26. The DC blocking capacitance 26 isconnected via bond wires 32 to the power supply connection layer 36.

The active die 24 is connected via bond wires 40 to the matching circuit30. The power supply connection layer 36 is connected with one side ofthe decoupling capacitance 38. The other side of the decouplingcapacitance 38 is connected to ground 34. The bond wire 28 and the DCblocking capacitance 26 forms the INSHIN-circuit. The DC blockingcapacitance 26 is a RF short This means that any circuitry may beconnected to the DC blocking capacitance 26 without effecting theoperation of the RF matching circuit 30. The active die 24 is connectedto the matching circuit 30 through bond wires 40.

By connecting the power supply connection layer 36 and the decouplingcapacitance 38 to the DC blocking capacitance 26, the path length fromthe decoupling capacitance 38 to the active die 24 is dramaticallyreduced. The result is a much higher decoupling frequency and a simpleconnection to the power supply connection layer 36.

In other words, the power supply connection layer 36 is connecteddirectly to the DC blocking capacitance 26 between the INSHIN inductorwhich is formed by the bond wire 28, and ground. The DC blockingcapacitance 26 is a very good short for the working frequency andconnections which are made to it, do not influence at all theperformance of the active die 24 or the matching circuit 30. Directlyconnecting the power supply connection layer 36 to the DC blockingcapacitance 26 will make the ¼ wavelength line 12 of FIG. 1 obsolete.

Also the decoupling capacitance 38 to short the unwanted differentialfrequencies or tones can be placed directly next to the active die 24.This is a dramatic decrease of path length from the decouplingcapacitance 38 to the active die 24, resulting in a good short forfrequencies up to 50 MHz.

FIG. 3 is a diagram showing the difference in magnitude of the impedancebetween an RF amplifier device with traditional decoupling shown ascurve A and a RF amplifier device with a decoupling arrangement of theinvention shown as curve B. The unit of the vertical axis is Ω. The unitof the horizontal axis is Hz. In both decoupling arrangements, simple100 nF 1206 SMD capacitances are used.

When an impedance level of 0.5 Ω is considered as a sufficient short,then the traditional decoupling works from 30 kHz to 20 MHz as curve Ais below the 0,5 Ω level up to 20 MHz. As curve B stays below to 0,5 Ωline up to 75 MHZ, the decoupling arrangement of the invention worksfrom 30 kHz to 75 MHz. This clearly shows the superior performance ofthe RF amplifier device having the decoupling arrangement of theinvention.

FIG. 4 shows in the upper part a side view and in the lower part a topview of a standard RF discrete power transistor. The standard powertransistor 42 is mounted on a mounting base 44. An aluminum nitrite(AlN) ring 46 is mounted between the transistor 42 and the mounting base44. The transistor 42 is connected to other circuit parts by the drainlead 48 and the gate lead 50.

In a standard transistor package used for LDMOS-base stationtransistors, a copper/TUNGSTEN alloy mounting base connects thetransistor 42 electrically and thermally to ground. On this mountingbase 44, an aluminum nitrite ring 46 is mounted, on which the leads 48,50 are connected, electrically separated from the mounting base 44 bythis AIN ring 46. The leads 48, 50 connect the transistor 42 to thematching network on the circuit board and transport both DC-biases aswell as RF power.

FIG. 5 shows the case of an adapted RF discrete power transistor withtwo extra leads 52, 54. In FIG. 5, equivalent parts with equivalentnumbers are shown as in FIG. 4. The adapted power transistor 42 of FIG.5 transports the RF power by the middle leads 48 and 50 and the DC biasto the transistor 42 of FIG. 5 as provided by one or both outer leads 52and 54. Furthermore, the leads 52 and 54 are used for decoupling thetransistor 52 of FIG. 5 from the connected circuit parts. The adapted RFpower transistor of FIG. 5 is used in the following descriptions ofINSHIN decoupling in discrete transistors.

FIGS. 6 to 9 show different embodiments of INSHIN decoupling and powersupply in discrete transistors. In the FIGS. 6 to 9 only one half ofdrain lead of a transistor is shown.

The transistor die 56 is connected to the INSHIN capacitance 64 by thebond wires 58, 60 and 62. The capacitance 64 and the inductance of thebond wires 58, 60 and 62 compensate the capacitance of an input oroutput terminal of the transistor die 56. The transistor die 56 isconnected to the RF lead 70 by the bond wires 66 and 68. The decouplingand power supply lead 76 is connected to the INSHIN capacitance 64 bythe bond wires 72 and 74. The decoupling and power supply lead 76provides power to the transistor die 56 and is used for decoupling ofthe transistor die 56 from the connected circuit parts. The decouplingand power supply lead 76 and the RF lead 70 are mounted on the AIN ring78.

The transistor die 80 is connected to the INSHIN capacitance 92 by thebond wires 82, 84 and 86. The function of the INSHIN capacitance 92 andthe bond wires 82, 84 and 86 is the same as described above. Thetransistor die 80 is connected to the RF lead 70 by the bond wires 88and 90. The INSHIN capacitance 92 is connected to the decoupling andpower supply lead 98 by the bond wires 94 and 96. The decoupling andpower supply lead 98 is also mounted on the AIN ring 78.

This embodiment is the basic embodiment. The embodiment shows a discretetransistor with an external connection to the INSHIN capacitance.Advantages of this embodiment are the simple construction, the supplywith DC bias and the decoupling through separate leads, no need for a ¼wavelength line on RF paths to bias the transistor and a biasing anddecoupling close to the transistor.

The embodiment of FIG. 7 is rather similar to the embodiment of FIG. 6.In FIG. 7 a decoupling capacitance 100 makes the difference to theembodiment of FIG. 6. The decoupling capacitance 100 is connected to thedecoupling in power supply lead 76 by the bond wires 106 and 108, andthe decoupling capacitance 100 is connected to the decoupling and powersupply lead 98 by the bond wires 102 and 104. The bond wires 58, 60 and62 connect the transistor die 56 to the INSHIN capacitance 64 and to thedecoupling capacitance 100. The bond wires 82, 84 and 86 connect thetransistor die 80 to the INSHIN capacitance 92 and to the decouplingcapacitance 100. The capacitance 100 could be made for instance as astrip of Hi-K material.

An advantage of the capacitance 100 is that the capacitance enhances thelow frequency decoupling significantly. Further advantages are that theDC bias and the decoupling of the transistor 56 and 80 is providedthrough separate leads 76, 98 and that there is no need for ¼ wavelengthline to bias the transistors 56 and 80.

FIG. 8 shows an embodiment, which is rather similar to the embodiment ofFIG. 7. The main difference between FIG. 7 and FIG. 8 is, that thedecoupling capacitance 100 of FIG. 7 is divided into two decouplingcapacitances 110 and 120. The decoupling capacitance 110 is connected tothe decoupling in power supply lead 76 by the bond wires 116 and 118.The decoupling capacitance 110 is connected to the INSHIN capacitance 64by the bond wires 112 and 114. The decoupling capacitance 120 isconnected to the INSHIN capacitance 92 by the bond wires 122 and 124.The decoupling capacitance 120 is connected to the decoupling and powersupply lead 98 by the bond wires 126 and 128.

The two decoupling capacitances 110 and 120 at the side of thetransistor dies 56 and 80 and at the side of the INSHIN capacitance dies64 and 92 allow shorter drain wires to the lead. This may beadvantageous for matching purposes.

FIG. 9 shows an embodiment, in which the transistor die 56 is connectedto a capacitance 130, which combines an INSHIN and a decouplingcapacitance by the bond wires 132, 134 and 136. The transistor die 80 isconnected to the capacitance 130 by the bond wires 138, 140 and 142. Thecapacitance 130 is connected to the decoupling and power supply lead 76by the bond wires 144 and 146. The capacitance 130 is connected to thedecoupling and power supply lead 98 by the bond wires 148 and 150.

If the decoupling capacitance is sufficient to effectively shorten thecarrier frequency, the INSHIN capacitance may be omitted to combine twofunctions in one capacitance. The first function which is combined isshortened of low frequency products, which is normally done by thedecoupling capacitance. The second function is shortening the signal atthe carrier frequency which is normally done by the INSHIN capacitance.

Advantages of the embodiment of FIG. 9 are saving of space inside thetransistor, saving the use of the INSHIN capacitance, very good lowfrequency decoupling. Further advantages of this embodiment is that itis much easier to manufacture than the embodiments of the state of theart and the short drain bond wires, which may be advantageous inimproving matching properties.

In order to demonstrate the special features of the embodiment shown inthe FIGS. 6–9, only the differences between the embodiments aredescribed. Equal parts have equal numbers.

FIGS. 10 to 13 show different embodiments of INSHIN decoupling and powersupply in a module. All designs used in a discrete transistor can alsobe used in a module. A module features a mounting base as a discretetransistor, on which a printed circuit board (pcb) is soldered. Thiscircuit board may contain all or a part of the matching network, all orsome parts of the bias circuit as well as decoupling capacitances. Anadvantage of a module is that a multilayer printed circuit board can beused (where a discrete transistor only has a AIN ring), which extendsthe number of connecting possibilities.

FIG. 10 shows in the upper part a top view of an embodiment of INSHINdecoupling and power supply in a module. A side view is shown in thelower part of FIG. 10.

The transistor die 210 is connected to a decoupling capacitance 208 bybond wires 218, 220 and 222. The transistor die 210 is also connected inparallel to a RF path 206 by bond wires 230 and 232. The decouplingcapacitance 208 is connected to a decoupling and power supply path 202.

A transistor die 212 is connected to the decoupling capacitance 208 bybond wires 224, 226 and 228. The transistor die 212 is connected to theRF path 206 by bond wires 234 and 236. The decoupling capacitance 208 isconnected to the decoupling and power supply path 204.

The lower part of FIG. 10 shows a side view of the embodiment. The RFpath 206 is mounted on a top layer 238 and the decoupling and powersupply path 204 is mounted on a middle layer 240. A connection is madeby vias to the top layer 238, where decoupling capacitances can beplaced.

The advantages of the embodiment are that DC biasing and decoupling isdone through separate paths on the printed circuit board 200. There isno need for ¼ wavelength line on the RF path 206 to bias the transistor210 and 212. The biasing and decoupling is close to the transistors 210and 212. There is no DC bias current to flow laterally throughcapacitance. Instead the DC bias current flows from the printed circuitboard 200 to the transistor dies 210, 212 via the bond wires 218, 220,222, 224, 226, 228, 230, 232, 234 and 236.

FIG. 11 shows basically the same as FIG. 10. The difference is that theRF signal path 206 is in the middle layer 240 and the decoupling andpower supply path 204 is on the top layer 238. The advantages are asdescribed above. A further advantage of this design is that is hasshorter drain bond wires. This is an advantage to the previous layoutfor broad band matching.

FIG. 12 shows an embodiment rather similar to the embodiments of FIGS.10 and 11. The difference to the embodiments described above is thatthere is a separate decoupling capacitance 208. The transistor die 210is connected to the INSHIN capacitance 214, the decoupling capacitance208 and to the decoupling and power supply path by the bond wire 242,244 and 246. The transistor die 210 is connected to the RF path 206 bythe bond wires 230 and 232. The transistor 212 is connected to theINSHIN capacitance 216, the decoupling capacitance 208 and thedecoupling and power supply path by the bondwires 248, 250 and 252. Thetransistor 212 is connected to the RF path 206 by the bondwires 234 and236. The lower part of FIG. 12 shows that the RF path 206 is on the toplayer 238 and the decoupling and power supply path 204 is on themiddlelayer 240.

The embodiment as shown in FIG. 12 is like the embodiment as shown inFIG. 7, but the power supply connection is in parallel to the dies 208,210, 212, 214 and 216.

Advantages are that the capacitance 208 inside the transistor enhancesthe low frequency decoupling dramatically. The biasing with DC and thedecoupling of the transistors 210 and 212 is done by separate paths onthe printed circuit board 200. There is no need for ¼ wavelength line tobias the transistors 210 and 212. The biasing and the decoupling isclose to the transistors 210 and 212. There is no bias current to flowlaterally through the capacitances 208, 214 and 216. Instead biascurrent flows from the printed circuit board to the transistors 210 and212 via bond wires 242, 244, 246, 248, 250 and 252.

FIG. 13 shows an embodiment of a combined INSHIN and decouplingcapacitance 266 as FIG. 9. The transistor 210 is connected to thedecoupling and power supply path and to the capacitance 266 by the bondwires 254, 256 and 258. The transistor 212 is connected to the combinedcapacitance 266 and to the decoupling and power supply path by the bondwires 260, 262 and 264. The transistor 210 is connected to the RF path206 by the bond wires 230 and 232. The transistor 212 is connected tothe RF path 206 by the bond wires 234 and 236. The lower part of FIG. 13shows the structure of the embodiment.

An advantage of the embodiment are that the embodiment saves spaceinside the module. Furthermore the embodiment saves the use of INSHINcapacitances, it has a very good low frequency decoupling, while at thesame time the manufacture is easier if no separate INSHIN capacitance isneeded and the drain bond wires are rather short. This is useful formatching.

New characteristics and advantages of the invention covered by thisdocument have been set forth in the foregoing description. It will beunderstood, however, that this disclosure is, in many respects, onlyillustrative. Changes may be made in details, particularly in matters ofshape, size, and arrangement of parts, without exceeding the scope ofthe invention. The scope of the invention is, of course, defined in thelanguage in which the appended claims are expressed.

1. An RF amplifier device (22) comprising an amplifier element (24) witha frequency dependent gain, said frequency dependence being caused by aninput and/or an output capacitance, said frequency dependence beingcompensated by a compensating circuit (26, 28), said compensatingcircuit (26, 28) for compensating for said frequency dependence beingdirectly connected to a power supply connection terminal (36), saidpower supply connection terminal (36) being connected to a decouplingcircuit (38).
 2. A device as claimed in claim 1, comprising an amplifierelement (24) and a compensating circuit (26, 28) comprising an internalshunt inductor having a compensating inductance (28) and a compensatingcapacitance (26), said compensating circuit being arranged in parallelto the amplifier element output (24) to compensate an input and/or anoutput capacitance of the device, said device further comprising a powersupply terminal (36) connected to said amplifier element through saiddecoupling circuit (38), whereby the decoupling circuit (38) is coupledto the compensating capacitance (26) via said power supply terminal(36).
 3. A device as claimed in claim 2, wherein the decoupling circuit(38) is coupled to the compensating capacitance (26) through aninductance element (32).
 4. A device as claimed in claim 2, wherein theinductance element comprises at least one bond wire (28, 32, 40).
 5. Adevice as claimed in claim 2, wherein the decoupling circuit (38) isconnected to ground (34) and the power supply (36).
 6. A device asclaimed in claim 3, wherein the decoupling circuit comprises at leastone decoupling capacitance (38).
 7. A device as claimed in claim 2,wherein a power supply line is provided comprising a power supplyconnection area, a decoupling circuit connection area and a bond wireconnection area, which line is arranged next to the compensatingcapacitance (26).
 8. A device as claimed in claim 2, wherein said RFamplifier device (24) is a transistor.
 9. A device as claimed in claim1, in which the amplifier element (24) with the compensating circuit(26, 28), the decoupling circuit and the connection line having theconnection area for a power supply (36) and a connection area for thedecoupling circuit (38) are arranged on a circuit board, wherein theconnection line is located on the circuit board next to the compensatingcapacitance (26).
 10. A device as claimed in claim 1, comprising anamplifier element (56, 80) and a compensating circuit comprising aninternal shunt inductor having a compensating inductance (58, 60, 62) inseries with a compensating capacitance (64, 92), which are arranged inparallel to a terminal of the amplifier element (56, 80) to compensate aterminal capacitance of the amplifier element (56, 80), said devicefurther comprising a decoupling and power supply lead (76, 98) connectedto the compensating capacitance (64, 92) and/or a decoupling circuit(100) and/or a combination of the compensating capacitance and thedecoupling circuit (130).
 11. A device as claimed in claim 10, whereinthe terminal of the amplifier element (56, 80) is an input terminaland/or an output terminal of the amplifier element (56, 80).
 12. Adevice as claimed in claim 10, wherein the decoupling circuit (100) isconnected to the compensating capacitance (64, 92) through an inductanceelement (58, 60, 62, 82, 84, 86).
 13. A device as claimed in claim 10,wherein the inductance element comprises at least one bond wire (58, 60,62, 82, 84, 86).
 14. A device as claimed in claim 10, wherein thedecoupling circuit (100, 110, 120) and/or the combination of thecompensating capacitance and the decoupling circuit (130) are/isconnected between the decoupling and power supply lead (76, 98) and thecompensating capacitance (64, 92) or between the decoupling and powersupply lead (76, 98) and the terminal of the amplifier element (56, 80).15. A device as claimed in claim 10, wherein the decoupling circuitcomprises at least one decoupling capacitance (100, 110, 120).
 16. Adevice as claimed in claim 10, wherein the amplifier element (56, 80) isa transistor.
 17. A device as claimed in claim 1, in which the amplifierelement (56, 80) with the compensating circuit and/or the decouplingcircuit (100, 110, 120) and/or the combination of the compensatingcapacitance and the decoupling circuit (130) and the decoupling andpower supply lead (76, 98) are arranged on a circuit board.
 18. A modulecomprising an RF amplifier device as claimed in claim 1 comprising amounting base (44) for a discrete transistor on which a printed circuitboard (pcb) is soldered; a matching network; a bias circuit; at leastone decoupling capacitance.
 19. A module as claimed in claim 18, whereinthe printed circuit board is a multilayer printed circuit board (200).20. A module as claimed in claim 18, wherein the printed circuit board(200) contains all or a part of the matching network and/or the biascircuit.
 21. A module as claimed in claim 18, wherein a signal path(206) is on a top layer (238) and a decoupling and power supply path(202, 204) is on a middle layer (240) of the pcb (200) or vice versa.22. A module as claimed in claim 18, wherein the decoupling and powersupply path (202, 204) is in parallel to the dies of an amplifierelement (210, 212) and/or a compensating capacitance (214, 216) and/or adecoupling circuit (208) and/or a combination of the compensatingcapacitance and the decoupling circuit (266).
 23. A method fordecoupling an RF amplifier device (22) comprising an amplifier element(24) with a frequency dependent gain, said frequency dependence beingcaused by an input and/or an output capacitance, said frequencydependence being compensated by a compensating circuit (26, 28), saidcompensating circuit (26, 28) for compensating for said frequencydependence being directly connected to a power supply connectionterminal (36), said power supply connection terminal (36) being coupledto ground (34) via a frequency dependent impedance (38).